Systems and methods for the dynamic range compression of multi-bearer single-carrier and multi-carrier waveforms

ABSTRACT

The present invention is related to methods and apparatus that can advantageously reduce a peak to average signal level exhibited by single or by multicarrier multibearer waveforms. Embodiments of the invention further advantageously can manipulate the statistics of the waveform without expanding the spectral bandwidth of the allocated channels. Embodiments of the invention can be applied to either multiple carrier or single carrier systems to constrain an output signal within predetermined peak to average bounds. Advantageously, the techniques can be used to enhance the utilization of existing multicarrier RF transmitters, including those found in third generation cellular base stations. However, the peak to average power level managing techniques disclosed herein can apply to any band-limited communication system and any type of modulation. The techniques can apply to multiple signals and can apply to a wide variety of modulation schemes or combinations thereof.

RELATED APPLICATIONS

This application is a continuation application of U.S. application Ser.No. 09/910,422, filed Jul. 20, 2001, which claims the benefit under 35U.S.C. § 119(e) of U.S. Provisional Application No. 60/220,018, filedJul. 21, 2000, the disclosures of which are hereby incorporated byreference in their entireties herein.

A co-pending and co-owned patent application with application Ser. No.09/910,477, filed on Jul. 20, 2001, commonly owned and filed on the sameday as the present application, is hereby incorporated herein in itsentirety by reference thereto.

BACKGROUND OF THE INVENTION

1. Field of the Invention

The present invention generally relates to electronics. In particular,the present invention relates to communications systems.

2. Description of the Related Art

The rapid commoditization of the cellular, personal communicationservice (PCS) and wireless industries has resulted in the emergence ofnew digital radio standards, which support the emergence of high userbandwidth requirements. For example, third generation (3G) digitalwide-band code division multiple access (W-CDMA) and Enhanced Data GSM(Group System for Mobile Communications) Environment (EDGE) airinterface standards exploit signal processing techniques that cangenerate radio and baseband waveforms with a relatively high peak powerto average power ratio.

The signals amplified by a wireless base station include multiplesignals, which are combined to a multi-bearer waveform. The number ofvoice and data connections represented within the multi-bearer waveformcan vary randomly and vary over time. Occasionally, the informationsources that are combined to form the multi-bearer waveform can co-alignand generate a relatively large instantaneous signal peak or crest. Inone example, the relatively large instantaneous signal peak is about 10times higher in power than a nominal or average output level.

In practice, the alignment that generates a relatively largeinstantaneous signal peak occurs with a relatively low probability.Despite the relatively low probability, however, the dynamic range ofthe entire signal processing chain of a base station should besufficient to handle the large instantaneous signal peak in order totransmit the signal without error.

One conventional approach is to design the base station to accommodatethe relatively rare, but large, signal peak. As a result, the basestation is significantly overdesigned, which results in a significantincrease to the cost of the base station. In particular, the cost andthe size of the radio frequency (RF) amplifier of the base station aredeleteriously affected. For example, such an approach disadvantageouslylowers the efficiency of the RF amplifier, as a higher powered RFamplifier will waste significantly larger amounts of power for biasesand the like. Further, the extra power dissipation is correspondinglydissipated with larger and more costly heat management techniques.

In addition, the relatively large dynamic range imposed upon the basestation by the relatively large signal peak typically requires that theupconversion circuitry, the digital to analog converters, the digitalsignal processing circuits, and the like also accommodate the relativelylarge dynamic range.

In another conventional approach, the signal waveform is hard limited toreduce the dynamic range of the relatively rare signal peaks. Thisallows a relatively lower power RF transmitter to be used to transmitthe signal, which allows the RF transmitter to operate with relativelylarger efficiency. However, conventional hard limiting techniques areimpractical because hard limiting generates distortion energy, whichcauses interference in adjacent channels.

SUMMARY OF THE INVENTION

Embodiments of the present invention include apparatus and methods thatovercome the disadvantages of the prior art by manipulating amultibearer waveform, which can include single carrier or multiplecarrier waveforms that reduce the peak to average ratio of themultibearer waveform. Advantageously, embodiments of the presentinvention allow radio frequency (RF) base stations to be more efficient,compact, and lower in cost than conventional base stations.

Embodiments of the invention permit significant reduction to the cost toprovision digital and analog signal processing chains in communicationsystems. Embodiments of the invention may be applied to a variety ofcommunications systems including both wire and wireless communicationssystems such as cellular, personal communications service (PCS), localmultipoint distribution systems (LMDS), and satellite systems.

One embodiment of the invention includes a predictive weight generatorthat reduces an amount of waveshaping processing applied to a pluralityof input symbol streams by a waveshaping circuit. The predictive weightgenerator includes pulse-shaping filter emulation circuits that receivethe plurality of input symbol streams. A pulse-shaping filter emulationcircuit can be constructed from a pulse-shaping circuit. The predictiveweight generator further includes mixers coupled to the pulse-shapingfilter emulation circuits and coupled to digital numerically controlledoscillators that upconvert actual outputs of actual pulse-shapingfilters for the input symbol streams. The outputs of the mixers aresummed by a summing circuit to simulate a composite signal and tothereby predict an amplitude of an actual composite signal. A comparatorcompares the predicted amplitude to a threshold level and providesweight value modifications to the waveshaping circuit in response to thecomparison in real time.

One embodiment of the invention includes a post-conditioning circuitthat generates a de-cresting pulse that can decrease an amplitude of asignal peak of a composite multicarrier signal in real time. Thecomposite multicarrier signal includes a plurality of input symbolstreams that are pulse-shaped and frequency upconverted to a pluralityof upconverted streams. The post-conditioning circuit includes acomparator, a weight generator, an impulse generator, a multipliercircuit, and a bandpass filter.

The comparator compares the composite multicarrier signal to apredetermined threshold such that the comparator activates an outputwhen the composite multicarrier signal exceeds the predeterminedthreshold. The weight generator receives the plurality of upconvertedstreams and phase information from a plurality of oscillators as inputs.The weight generator also receives carrier waveforms for the pluralityof upconverted streams so that the weight generator can determine anupconverted stream's contribution to the composite multicarrier signal'ssignal peak. The weight generator calculates a weight value for theupconverted stream approximately proportionately to the upconvertedstream's contribution to the composite multicarrier signal's signalpeak.

The impulse generator provides an impulse as an output in response tothe output of the comparator. The impulse generator also controls aduration of the generated impulse in response to the output of thecomparator. The multiplier circuit multiplies the weight value from theweight generator with the impulse from the impulse generator to generatea scaled impulse. The bandpass filter filters the scaled impulse to afrequency band that corresponds to the upconverted stream's allocatedfrequency band to generate the de-cresting pulse.

In one embodiment, multiple pulses are injected to de-crest thecomposite multicarrier signal. The multiple pulses can advantageouslyprevent the injection of signal energy to unutilized adjacent channelallocations.

One embodiment of the invention includes a pulse-shaping circuit thatreduces a probability of an alignment in amplitude and phase of similarsymbols in a plurality of input symbol streams. The plurality of inputsymbol streams are eventually upconverted and combined to a compositedata stream and include at least a first input symbol stream and asecond input symbol stream. Advantageously, a reduction in theprobability of the alignment reduces a probability of a large signalcrest in the composite data stream.

The pulse-shaping circuit includes a plurality of pulse-shaping filters,which pulse-shape the plurality of input symbol streams to acorresponding plurality of baseband streams. The pulse shaping circuitfurther includes a plurality of multipliers, which upconvert theplurality of baseband streams to a plurality of upconverted streams, anda summing circuit that combines the upconverted streams to the compositesignal. The pulse shaping circuit also includes a delay circuit in atleast a first data path. The first data path is a path from an inputsymbol stream to the composite data stream. The delay circuit delaysdata in the first data path by a fraction of a symbol period relative todata in a second data path to stagger symbols in the symbol streams.

One embodiment of the invention includes a composite waveformde-cresting circuit that digitally generates at least one de-crestingphase shift in real time that allows a composite multicarrier signal tobe generated with a decrease in an amplitude of a signal peak.Advantageously, the circuit decreases the amplitude of the signal peakof the composite multicarrier signal without altering an amplitude ofthe plurality of input symbol streams. The circuit includes acomputation circuit, a comparator, at least one impulse generator, andat least one phase shifter.

The computation circuit receives the plurality of upconverted streamsand a phase information from a plurality of oscillators that providecarrier waveforms for the plurality of upconverted streams. Thecomputation circuit predicts a level in the composite multicarriersignal. The comparator compares the predicted level of the compositemulticarrier signal from the computation circuit to a predeterminedthreshold and the comparator activates an output when the compositemulticarrier signal exceeds the predetermined threshold.

The weight generator receives the plurality of upconverted streams and aphase information from the plurality of oscillators that provide carrierwaveforms for the plurality of upconverted streams. The weight generatorcalculates a weight value for an upconverted stream in the plurality ofupconverted streams, where the weight value is approximatelyproportional to the upconverted stream's contribution to the predictedlevel of the composite multicarrier signal's signal peak.

The impulse generator provides an impulse as an output in response tothe output of the comparator. The impulse generator also controls aduration of the generated impulse in response to the output of thecomparator. The multiplier circuit multiplies the weight value from theweight generator with the impulse from the impulse generator to generatea scaled impulse. The bandpass filter that filters the scaled impulse toa frequency band that corresponds to the upconverted stream's allocatedfrequency band to generate a de-cresting phase-shift control signal. Thephase shifter modulates a relative phase of the upconverted stream inresponse to the de-cresting phase-shift control signal.

BRIEF DESCRIPTION OF THE DRAWINGS

These and other features of the invention will now be described withreference to the drawings summarized below. These drawings and theassociated description are provided to illustrate preferred embodimentsof the invention and are not intended to limit the scope of theinvention.

FIG. 1 illustrates a waveshaping circuit according to one embodiment ofthe present invention.

FIG. 2 illustrates a complementary cumulative distribution function(CCDF) curve for an intrinsic W-CDMA multicarrier signal.

FIG. 3 illustrates a multi-carrier waveshaping circuit according to oneembodiment of the present invention.

FIG. 4 illustrates a waveshaping circuit according to an embodiment ofthe present invention that adaptively modifies the waveshapingprocessing to fit predetermined criteria.

FIG. 5 illustrates a preconditioning circuit according to an embodimentof the present invention.

FIGS. 6A-E illustrate an example of the operation of the preconditioningcircuit shown in FIG. 5.

FIG. 7 graphically represents limiting with a relatively soft signallevel threshold and limiting with a relatively hard signal levelthreshold.

FIG. 8 illustrates another preconditioning circuit according to anembodiment of the present invention.

FIG. 9 illustrates a waveshaping circuit according to an embodiment ofthe present invention.

FIG. 10 consists of FIGS. 10A and 10B and illustrates a multicarrierde-cresting circuit according to an embodiment of the present invention.

FIGS. 11A-E illustrate an example of the operation of the multicarrierde-cresting circuit shown in FIG. 10.

FIGS. 12A-C are power spectral density (PSD) plots of de-cresting with asingle Gaussian pulse.

FIGS. 13A-E illustrate de-cresting with multiple Gaussian pulses.

FIGS. 14A and 14B illustrate the results of a complementary frequencydomain analysis of a multicarrier de-cresting circuit.

FIG. 15 illustrates one embodiment of a de-cresting pulse generationcircuit.

FIG. 16 illustrates a pulse-shaping filter according to an embodiment ofthe present invention.

FIG. 17 consists of FIGS. 17A and 17B illustrates a phase-modulatingwaveshaping circuit according to an embodiment of the present invention.

DETAILED DESCRIPTION OF PREFERRED EMBODIMENTS

Although this invention will be described in terms of certain preferredembodiments, other embodiments that are apparent to those of ordinaryskill in the art, including embodiments which do not provide all of thebenefits and features set forth herein, are also within the scope ofthis invention. Accordingly, the scope of the present invention isdefined only by reference to the appended claims.

FIG. 1 illustrates a waveshaping circuit 100 according to one embodimentof the present invention. A waveshaping circuit can be adapted to shapeeither single data streams or multiple input streams with multiplebaseband signals. The waveshaping circuit 100 shown in FIG. 1 is adaptedto shape a single input data stream to a single shaped output datastream. Other embodiments that are adapted to shape and to combinemultiple input signals to a shaped output data stream are describedlater in connection with FIGS. 3, 4, 9, 10, 15, 16, and 17.

An input symbol stream 102 is applied as an input to the waveshapingcircuit 100. The input symbol stream 102 can include data for cellulartelephone communications, data communications, and the like. Thewaveshaping circuit 100 generates an output sample stream 104 as anoutput. Advantageously, the output of the waveshaping circuit 100 has alower dynamic range than the input symbol stream 102. The lower dynamicrange of the output sample stream 104 allows a base station to processand to amplify the output sample stream 104 with lower power and lowerdynamic range components.

The waveshaping circuit 100 includes a preconditioning stage 106, apulse-shaping and frequency translating circuit 108, and apost-conditioning circuit 110. The waveshaping circuit 100 can replacean upconversion circuit or portions of the waveshaping circuit 100 canbe used to supplement existing upconversion circuits.

The preconditioning stage 106 includes a preconditioning circuit 112. Inalternate embodiments, where multiple input baseband signals are shapedand combined, the preconditioning stage 106 can include multiplepreconditioning circuits. The preconditioning circuit 112 appliesnonlinear processing to the input symbol stream 102 on a symbol bysymbol basis. In one embodiment, the preconditioning circuit 112 appliesa soft nonlinear compression function, which severely compressesrelatively extensive signal peaks and compresses relatively modestsignal peaks into a predefined signal range. The output of thepreconditioning circuit 112 is provided as an input to the pulse-shapingand frequency translating circuit 108. At this point in the data flow,bandwidth expansion is not a concern since the output of thepreconditioning circuit 112 exhibits a white spectral characteristic.Further details of the preconditioning circuit 112 are described laterin connection with FIGS. 5, 6, 7, and 8.

The illustrated pulse-shaping and frequency translating circuit 108includes a pulse-shaping filter 114, a digital numerically controlledoscillator (NCO) 116, and a mixer 118. The pulse-shaping filter 114 mapsthe source bits of the output of the preconditioning circuit 112 to abaseband pulse. The output of the pulse-shaping filter 114 and an outputof the digital NCO 116 are applied as inputs to the mixer 118. In oneembodiment of the waveshaping circuit 100, the pulse-shaping andfrequency translating circuit 108 is implemented with conventionalcomponents.

In a conventional base station without waveshaping, a sequence of inputmodulation symbols is streamed into a pulse-shaping filter and to afrequency upconversion circuit. The modulation symbols usually exhibit awhite frequency spectral density and it is not until the symbol rate isstepped up to the higher sample stream rate by the pulse-shaping filterthat the new modulation sample stream is band-limited by the actions ofthe filter. The baseband sample stream output of the pulse-shapingfilter can be shifted to a new digital carrier frequency bymultiplication with the output of the digital NCO. The input symbolstream 102 often is a composite of many symbol streams drawn from anumber of active voice and data users. Consequently, on occasion, thesesymbol streams linearly (vectorially) add up to a relatively largesignal peak when relatively many users simultaneously transmit a similaror identical modulation symbol.

The mere preconditioning of the input symbol stream 102 by thepreconditioning circuit 112 does not adequately reduce peaks in theoutput of the mixer 118 due to Gibbs-type phenomena in the pulse-shapingfilter 114. The Gibbs-type phenomena re-introduces signal peaks to thesignal stream as a natural consequence of filtering.

In order to compensate for the signal peaks from the pulse-shapingfilter 114, the waveshaping circuit 100 includes the post-conditioningcircuit 110. The post-conditioning circuit 110 includes a pulsegenerator 120 and a summing circuit 122. The pulse generator 120 detectssignal peaks and introduces via the summing circuit 122 a band-limitedGaussian pulse that destructively interferes with peaks in the output ofthe mixer 118 to reduce the peaks in the output sample stream 104.Although the destructive interference can temporarily undermine thewaveform integrity of the output sample stream 104, thepost-conditioning circuit 110 advantageously limits the upper peakvalues of the output sample stream 104 to a relatively precise dynamicrange.

This transitory degradation in the integrity of the output sample stream104 is tolerable, particularly in CDMA systems, because the introducederror energy is not de-spread in the signal recovery processingundertaken by the receiver. In one embodiment, the pulse generator 120generates a Gaussian pulse or a family of Gaussian pulses todestructively interfere with the signal peaks in the output of the mixer118. Advantageously, the error energy of a Gaussian pulse or family ofGaussian pulses is equally spread among W-CDMA spreading codes. Inaddition to their spectral characteristic, Gaussian pulses can begenerated relatively easily and with relatively low latency. In otherembodiments, the pulse generator 120 uses other types of band-limitedpulse shapes such as Blackman pulses, Hamming pulses, Square Root RaisedCosine (SRRC) pulses, Raised Cosine (RC) pulses, Sinc pulses and thelike to destructively interfere with and reduce the signal peaks.Further details of the post-conditioning circuit 110 are described laterin connection with FIGS. 10 to 17.

FIG. 2 illustrates a complementary cumulative distribution function(CCDF) curve for an intrinsic W-CDMA multicarrier signal. The W-CDMAmulticarrier signal is a multi-bearer waveform that includes a timevariant random number of data and voice connections which, on relativelyrare occasions, can co-align and generate a relatively largeinstantaneous signal peak. Although the relatively high amplitude signalpeaks are relatively rare, the probability of the occurrence of therelatively high amplitude signal peaks is non-zero and should beaccommodated by RF transmitters, base stations, and the like.

A horizontal axis 202 indicates output power relative to an average ormean power at 0 decibels (dB). A vertical axis 204 indicates the inverseprobability (1−P) of the CCDF curve. The curves in FIG. 2 illustrate anexample of the effects of peak power reduction by the destructiveinterference of a waveshaping circuit according to an embodiment of thepresent invention.

A first curve 206 corresponds to a typical, i.e., without waveshapingprocessing, CCDF curve with 10 dB of input back-off (ibo) for anintrinsic W-CDMA multicarrier signal. The first curve 206 illustratesthat without waveshaping processing, signal levels that exceed 5 dBabove the average signal level occur with a non-zero probability.Although the probability of such signal peaks is relatively low, theentire transmitter, which includes digital processors, analogupconverters, and power amplifiers, should accommodate such signalpeaks.

A second curve 208 illustrates an example of the effects of waveshapingprocessing according to an embodiment of the present invention. Thesecond curve 208 corresponds to a CCDF curve, where output signal peakshave been reduced through destructive interference by a waveshapingcircuit to limit the signal peaks to a selected threshold. In the secondcurve 208, the selected threshold is about 5 dB above the mean power.The selected threshold can be varied to correspond to a broad range ofvalues. In one embodiment, the selected threshold is fixed in a waveformshaping circuit. In another embodiment, a waveform shaping circuitmonitors the incoming data sequences and adaptively adjusts thecircuit's behavior to match with predetermined criteria. The reductionin signal peaks provided by embodiments of the present inventionadvantageously allows signals to be transmitted with more efficiency andwith lower power and lower cost RF amplifiers.

FIG. 3 illustrates a multi-carrier waveshaping circuit 300 according toone embodiment of the present invention, where the multi-carrierwaveshaping circuit 300 is adapted to reduce relatively high amplitudesignal peaks in a multi-carrier W-CDMA application. It will beunderstood by one of ordinary skill in the art that the number ofcarriers can vary over a broad range. The illustrated multi-carrierwaveshaping circuit 300 of FIG. 3 is shown with 3 carriers.

The multi-carrier waveshaping circuit 300 receives a first input symbolstream 302, a second input symbol stream 304 and a third input symbolstream 306 as inputs. The multi-carrier waveshaping circuit 300generates an output sample stream 308 by pulse-shaping, upconverting,combining, and waveshaping the input symbol streams.

The multi-carrier waveshaping circuit 300 includes a firstpreconditioning circuit 310, a second preconditioning circuit 312, athird preconditioning circuit 314, a first pulse-shaping filter 316, asecond pulse-shaping filter 318, a third pulse-shaping filter 320, afirst mixer 322, a second mixer 324, a third mixer 326, a first digitalnumerically controlled oscillator (NCO) 328, a second digital NCO 330, athird digital NCO 332, a post-conditioning pulse generator 348, a firstsumming circuit 350, a delay circuit 352, and a second summing circuit354.

The first preconditioning circuit 310, the second preconditioningcircuit 312, and the third preconditioning circuit 314 receive as inputsand process the first input symbol stream 302, the second input symbolstream 304 and the third input symbol stream 306, respectively, suchthat the peak to average ratio of each independent baseband inputchannel stream of modulation symbols is constrained within an initiallevel. One embodiment of a preconditioning circuit according to thepresent invention is described in greater detail later in connectionwith FIGS. 5 and 8.

The outputs of the first preconditioning circuit 310, the secondpreconditioning circuit 312, and the third preconditioning circuit 314,are applied as inputs to the first pulse-shaping filter 316, the secondpulse-shaping filter 318, and the third pulse-shaping filter 320,respectively, which map the inputs to baseband symbol streams.

The baseband symbol streams are applied as inputs to the first mixer322, the second mixer 324, and the third mixer 326. The first mixer 322,the second mixer 324, and the third mixer 326 mix the symbol streamswith a first output 340, a second output 342, and a third output 344 ofthe first digital NCO 328, the second digital NCO 330, and the thirddigital NCO 332, respectively, to upconvert and to produce multiplestreams of modulated channels. An output 334 of the first mixer 322, anoutput 336 of the second mixer 324, and an output 338 of the third mixer326 are combined to a composite signal by the first summing circuit 350.In addition, the outputs 334, 336, 338 constructively interfere anddestructively interfere with each other when combined. The constructiveinterference and the destructive interference can occur even where thesignals that are combined are individually pre-compensated to limithigh-amplitude signal peaks. As a result, the composite signal exhibitsan even greater dynamic range with a significantly greater peak toaverage power ratio than a single modulated channel.

Embodiments of the present invention advantageously compensate for therelatively high-amplitude signal peaks in composite signals caused byconstructive interference. In addition, embodiments of the presentinvention compensate for the relatively high-amplitude signal peaks withrelatively little, if any, injection of signal energy to adjacentchannel allocations. One embodiment that further advantageously detectsdestructive interference to at least partially disable thepre-compensation and the post-compensation applied to the input signalsand to the composite signal is described later in connection with FIG.9.

The post-conditioning pulse generator 348 compensates for the relativelyhigh-amplitude signal peaks in the composite signal by generatingmultiple Gaussian pulses, which are selected to destructively interferewith relatively high-amplitude signal peaks in the composite signal. Thepost-conditioning pulse generator 348 receives as inputs the outputs334, 336, 338 and analyzes the phase, frequency and amplitude of eachrespective channel carrier stream. This information permits the Gaussianpulse generator control to independently weigh a family of Gaussianpulses and to generate individual Gaussian pulses for each channelcarrier stream, where each pulse is centered at the respective carrierfrequency with a phase and amplitude selected to proportionally cancelthe particular channel's contribution to the instantaneous compositesignal's peak. The approach of utilizing multiple pulses is advantageousbecause signal energy is not injected into non-utilized adjacent channelallocations. Injection of signal energy to non-utilized adjacent channelallocations can undesirably interfere with other transmitters andsystems. Further details of the post-conditioning pulse generator 348are described later in connection with FIGS. 10-17.

The family of Gaussian pulses generated by the post-conditioning pulsegenerator 348 is applied as an input to the second summing circuit 354.The second summing circuit 354 sums the family of Gaussian pulses withan output of the delay circuit 352. The delay circuit 352 delays thecomposite signal from the first summing circuit 350 to align thecomposite signal with the Gaussian pulses generated by thepost-conditioning pulse generator 348. In one embodiment, the delaycircuit 352 delays the composite signal by the latency time associatedwith the post-conditioning pulse generator 348 minus the latency timeassociated with the first summing circuit 350. The delay circuit 352 canbe implemented with cascaded flip-flops, delay lines, and the like. Thesecond summing circuit 354 generates the output sample stream 308 as anoutput.

Waveshaping according to one embodiment of the present inventionincludes three processes: input preconditioning, pulse-shaping, andpost-conditioning de-cresting. Although each process can be configuredto operate independently within a waveshaping circuit, the operatingparameters for each process are preferably selected to complement eachother so that the waveshaping circuit as a whole functions optimally. Inone embodiment, the operating parameters are selected a priori andremain static. In another embodiment, a global de-cresting controlselects operating parameters adaptively and can adjust the operatingparameters dynamically.

FIG. 4 illustrates a waveshaping circuit 400 according to an embodimentof the present invention that adaptively modifies the waveshapingprocessing to fit predetermined criteria. It will be understood by oneof ordinary skill in the art that the number of individual input symbolstreams processed by the waveshaping circuit 400 can vary over a broadrange. The waveshaping circuit 400 shown in FIG. 4 is configured toprocess three such input symbol streams, which are a first input symbolstream 402, a second input symbol stream 404, and a third input symbolstream 406. As an output, the waveshaping circuit 400 generates anoutput sample stream 408.

The output sample stream 408 is advantageously monitored by ade-cresting control 416, which calculates and provides updates for thewaveshaping circuit 400 to allow the waveshaping circuit to adapt thewaveshaping processing to the input symbol stream. The de-crestingcontrol 416 also monitors the first input symbol stream 402, the secondinput symbol stream 404, and the third input symbol stream 406. Inaddition, the de-cresting control 416 receives a reference information418.

In response to the monitored input symbol streams 402, 404, 406, themonitored output sample stream 408, and the reference information 418,the de-cresting control 416 generates and provides parameter updates tothe first preconditioning circuit 410, to the second preconditioningcircuit 412, to the third preconditioning circuit 414, and to thepost-conditioning pulse generator 428. The parameter updates can includeupdates to coefficients used in digital filters, such as a finiteimpulse response (FIR) filter.

The first preconditioning circuit 410, the second preconditioningcircuit 412, the third preconditioning circuit 414, and thepost-conditioning pulse generator 428 shown in FIG. 4 are similar to thefirst preconditioning circuit 310, the second preconditioning circuit312, the third preconditioning circuit 314, and the post-conditioningpulse generator 348 described earlier in connection with FIG. 3. Furtherdetails of a preconditioning circuit are described later in connectionwith FIGS. 5, 7, and 8.

In one embodiment, the reference information 418 controls an amount ofdynamic range compression by the waveshaping circuit 400. The referenceinformation 418 can also be used to control a relative “hardness” orrelative “softness” of limiting as described later in connection withFIG. 7. The de-cresting control 416 permits the overall performance ofthe waveshaping circuit 400 to be monitored and permits adjustments tobe made to the parameters of individual, multiple or all of thesub-components of the waveshaping circuit 400. For example, thede-cresting control 416 can be used to adapt the processing of awaveshaping circuit to RF transmitters with a broad range of outputpower.

The de-cresting control 416 does not have to provide parameter updatesin real time. In one embodiment, the de-cresting control 416 isimplemented by firmware in a general purpose DSP or by a general-purposemicroprocessor or microcontroller. In one embodiment, the generalpurpose DSP or the general purpose microprocessor resides in an externalcircuit and interfaces to the first preconditioning circuit 410, to thesecond preconditioning circuit 412, to the third preconditioning circuit414, and to the post-conditioning pulse generator 428. In anotherembodiment, the de-cresting control 416, together with other componentsof the waveshaping circuit 400, is implemented with an applicationspecific integrated circuit (ASIC) or with a field programmable gatearray (FPGA).

FIG. 5 illustrates a preconditioning circuit 500 according to anembodiment of the present invention. The preconditioning circuit 500exploits the white spectral properties of an input symbol stream 502.The input symbol stream 502 includes a sequence of modulation symbolimpulses or rectangular pulses and occupies a relatively wide frequencyspectrum prior to pulse shaping by a pulse-shaping circuit. Thesubsequent pulse-shaping circuit filters a modified symbol stream 504and provides the overall spectral shaping to apply the specifiedbandwidth constraints.

One embodiment of the preconditioning circuit 500 advantageouslyexploits the pulse shaping by the pulse-shaping circuit to modify theoverall signal characteristics of the input symbol stream 502 byapplication of both linear and non-linear signal processing techniques.The spectral expansion induced by non-linear signal processing is laterremoved by the pulse-shaping circuit. In one embodiment, a subsequentpost-conditioning circuit, such as a post-conditioning pulse generator,is not permitted to process in a manner that would expand the spectrumoccupied by the processed signal. One embodiment of thepost-conditioning circuit accordingly processes the applied signal withlinear signal processing. However, exceptions are conceivable.

One embodiment of the preconditioning circuit 500 uses a pseudo randomsequence of pulses that is weighted to destructively interfere withselected pulses of the input symbol stream 502 and to select an amountof destructive interference.

With reference to FIG. 5, the illustrated preconditioning circuit 500includes a comparator 506, a first delay circuit 508, a weight generator512, a pseudo random sequence generator 514, a second delay circuit 516,a multiplier 518, and a summing circuit 520. Further operational detailsof the preconditioning circuit 500 are also described later inconnection with FIGS. 6A-E.

The input symbol stream 502 is applied as an input to the comparator 506and to the first delay circuit 508. The comparator 506 detects the levelof the instantaneous magnitude of the input symbol stream 502 andcompares the level to a reference level information 510 to determinewhether to apply signal preconditioning to the input symbol stream. Thereference level information 510 can be used to indicate a threshold or alimit to the magnitude and/or phase of a signal peak. In one embodiment,the reference level information 510 is statically predetermined a prioriand hard coded into the preconditioning circuit 500. In anotherembodiment, the reference level information 510 is adaptively providedby the de-cresting control, which can be an internal function or circuitof the waveshaping circuit or provided by a function or circuit externalto the waveshaping circuit. When the comparison indicates that signalpreconditioning is to be applied, the comparator 506 applies acorrection vector as an input to the weight generator 512.

The weight generator 512 receives the correction vector from thecomparator 506 and a pseudo random sequence from the pseudo randomsequence generator 514. In response to the correction vector and thepseudo random sequence, the weight generator 512 computes a weightfactor, which is applied as an input to the multiplier 518. The weightfactor, when applied to the pseudo random sequence, generates theappropriate correction vector that is linearly added to a delayedversion of the input symbol stream 502 to destructively interfere withrelatively high-amplitude signal peaks in the input symbol stream 502.In one embodiment, the weight factor is a scalar quantity that dependson a complex value of the input symbol stream 502 and a complex value ofthe pseudo random sequence.

The second delay circuit 516 delays the pseudo random sequence from thepseudo random sequence generator 514 to align the pseudo random sequencewith the weight factor from the weight generator. The weight factor andthe delayed pseudo random sequence are multiplied together by themultiplier 518 to generate the correction impulses.

The input symbol stream 502 is delayed by the first delay circuit 508.The first delay circuit 508 is configured to delay the input symbolstream 502 such that the input symbol stream 502 aligns with thecorrection impulses. In one embodiment, the first delay circuit 508delays the input symbol stream 502 by an amount of time approximatelyequal to the latency of the comparator 506, the weight generator 512,and the multiplier 518. The delays provide the preconditioning circuit500 with time to determine whether a modifying impulse or pulse is to beintroduced into the data flow in order to reduce a relatively highsignal peak or crest in the data sequence and to determine an amount ofa reduction in the magnitude and/or phase of the crest.

The delayed input symbol stream from the first delay circuit 508 islinearly summed by the summing circuit 520 with the correction impulsesfrom the multiplier 518. The linear superposition of the summing circuit520 generates the modified symbol stream 504 as an output. Therelatively high signal peaks in the input symbol stream 502 are reducedin the modified symbol stream 504 by destructive interference of theinput symbol stream 502 with the correction impulses.

Advantageously, the illustrated preconditioning circuit 500 can produceboth phase variations and amplitude variations in the input symbolstream 502 to de-crest the input symbol stream 502. The ability toprovide a phase variation finds particular utility in multi-carrierapplications, as will be described in connection with FIGS. 3, 4, 9, and10.

FIGS. 6A-E illustrate an example of the operation of the preconditioningcircuit 500 illustrated in FIG. 5. For clarity, the example shown inFIGS. 6A-E is drawn with the input symbol stream 502 and the pseudorandom sequence represented as scalar quantities. It will be understoodby one of ordinary skill in the art that both the input symbol stream502 and the pseudo random sequence are generally complex quantities withboth magnitude and phase. Also for clarity, the example shown in FIGS.6A-E does not show the delay in the first delay circuit 508 and in thesecond delay circuit 516.

In FIGS. 6A-E, a plurality of horizontal axes 602, 604, 606, 608, 610indicate time. FIG. 6A illustrates an example of the input symbol stream502, which is applied as an input to the preconditioning circuit 500.Dashed lines 612, 614 indicate a predetermined threshold level. Forexample, the predetermined threshold level can correspond to a peakoutput power level of an associated RF transmitter. In the example, fourevents 616, 618, 620, 622 exceed the predetermined threshold level.

FIG. 6B illustrates a time aligned pseudo random sequence of constantamplitude signal pulses from the pseudo random sequence generator 514.FIG. 6C illustrates a sequence of weight factors that are calculated bythe weight generator 512. The weight factors are applied to the pseudorandom sequence to generate the correction impulses. FIG. 6D illustratesa sequence of the correction impulses for the preconditioning circuit500.

FIG. 6E illustrates the modified symbol stream 504. The modified symbolstream 504 is the time-aligned linear superposition of the input symbolstream 502 with the correction impulses. The correction impulsesdestructively interfere with the four events 616, 618, 620, 622 shown inFIG. 6A so that an output level of the modified symbol stream 504 shownin FIG. 6E remains at or below the predetermined threshold level asshown by the dashed lines 612, 614. In one embodiment, thepreconditioning circuit 500 applies correction impulses to the inputsymbol stream 502 such that the modified symbol stream 504 does nottransgress beyond a selected signal level threshold.

FIG. 7 graphically represents limiting with a relatively soft signallevel threshold and limiting with a relatively hard signal levelthreshold. A horizontal axis 702 indicates an input level. A verticalaxis 704 indicates an output level.

A first trace 706 corresponds to limiting with a relatively hard signallevel threshold. In practice, the use of a single hard signal levelthreshold is not appropriate because the resulting complementarycumulative distribution function (CCDF) of the signal, as describedearlier in connection with FIG. 2, will not exhibit a smooth transitionbut rather an abrupt or rapid “cliff.” Such an approach often results inan unacceptably high error rate in the downstream receiver.

The preconditioning circuits according to the present inventionadvantageously overcome the disadvantages of relatively hard signallevel thresholding by employing a nonlinear weighting function thatprovides a varying amount of correction depending upon the magnitude ofthe input data stream. A second trace 708, a third trace 710, and afourth trace 712 represent exemplary transfer functions associated witha relatively soft signal-leveling threshold.

This approach of soft weighting eliminates the rapid onset of a “cliff”in the CCDF and replaces the abrupt cliff with a relatively soft regionin which the probability of a signal level exceeding a predeterminedsignal level is significantly less than that exhibited by the intrinsicinput symbol stream. At relatively high signal levels, the non-linearweighting function approaches a hard threshold, and a delay “cliff” inthe signal's CCDF occurs. The soft weighting approach does, however,provide a significant decrease in the level of error energy observed bythe downstream receivers.

The preconditioning circuit 500 operates by deliberately manipulatingthe amplitude and phase probability density function of the input signalwaveform so that the peak to average of the input signal's impulsestream is significantly lower than the original input waveform. Inpractice, any function or non-linear equation that exhibits behaviorwhich incurs desirable changes in the weight calculation can be employedby the preconditioning circuit 500. In one embodiment, the non-linearweighting function is expressed by Equation 1. In addition, thedeliberate insertion of Amplitude Modulation (AM), Phase Modulation(PM), or both can require an alternative function.

Equation 1 defines a family of soft preconditioning weighting functions.Equation 1 includes parameters α and β, which correspond to the degreeof non-linearity invoked. $\begin{matrix}{{V_{m}(t)} = {\frac{{V_{m}(t)}}{\left( {1 + \left( \frac{{V_{m}(t)}}{\beta} \right)^{\alpha}} \right)^{1/\alpha}}{\mathbb{e}}^{j{({\arg{({V_{m}{(t)}})}})}}}} & {{Eq}.\quad 1}\end{matrix}$

As α increases, the gain of the function increases, which permits anoverall level of preconditioning to be defined. Manipulation of βpermits the rate at which a hard clipping level is set.

FIG. 8 illustrates another preconditioning circuit 800 according to anembodiment of the present invention. The illustrated preconditioningcircuit 800 uses multipliers and coefficients to calculate a Taylorseries expansion of the non-linear weighting function shown in Equation1.

The approximation of the non-linear weighting function by the Taylorseries expansion includes at least three engineering compromises: delaylatency, power consumption, and precision of the Taylor seriesapproximation. The delay latency of the preconditioning circuit 800increases as a function of the order of the Taylor series expansion,i.e., increases with the number of multiplier stages. The powerconsumption of the preconditioning circuit 800 increases as the numberof multipliers is increased. The weighting function is less closelyapproximated by the Taylor series expansion, where fewer terms of theTaylor series expansion are computed.

The Taylor series approximation approach uses relatively extensive delaybalancing between each of the signal processing paths to ensure that thecalculated preconditioning function, represented in FIG. 8 as “p,”applies to the appropriate input samples. The illustratedpreconditioning circuit 800 computes the Taylor series expansion to thefourth order. It will be understood by one of ordinary skill in the artthat the preconditioning circuit 800 can be implemented in software aswell as in hardware.

The illustrated preconditioning circuit 800 includes a magnitudecomputation circuit 802, a first delay circuit 804, a first multiplier806, a second multiplier 808, a third multiplier 810, a second delaycircuit 812, a third delay circuit 814, a fourth delay circuit 816, afifth delay circuit 818, a sixth delay circuit 820, a coefficient bank822, a fourth multiplier 824, a fifth multiplier 826, a sixth multiplier828, a seventh multiplier 830, a summing circuit 832, an eighthmultiplier 834, and a ninth multiplier 836.

Generally, the input symbol stream is complex, with both an in-phasecomponent and a quadrature-phase component. The in-phase component ofthe input symbol stream, I_(input), is applied as an input to themagnitude computation circuit 802 and to the first delay circuit 804.The quadrature phase component of the input symbol stream, Q_(input), isapplied as an input to the magnitude computation circuit 802 and to thefirst delay circuit 804. The magnitude computation circuit 802 computesthe magnitude of the input symbol stream. In one embodiment, thecomputed magnitude corresponds approximately to a sum of squares.

An output of the magnitude computation circuit 802, termed “magnitude,”is applied as an input to the first multiplier 806, the second delaycircuit 812, and the fourth delay circuit 816. The first multiplier 806multiplies the magnitude by itself to produce a square of the magnitudeas an output. The output of the first multiplier 806 is applied as aninput to the second multiplier 808 and to the fifth delay circuit 818.

The second multiplier 808 receives and multiplies the output of thefirst multiplier 806 and an output of the second delay circuit 812. Thesecond delay circuit 812 delays the magnitude or the output of themagnitude computation circuit 802 by a latency associated with the firstmultiplier 806. The second multiplier 808 multiplies the squaredmagnitude from the first multiplier 806 with the first delayed magnitudefrom the second delay circuit 812 to generate a cubed magnitude.

The cubed magnitude output of the second multiplier is applied as aninput to the third multiplier 810 and to the sixth delay circuit 820.The first delayed magnitude output of the second delay circuit 812 isapplied as an input to the third delay circuit 814, which generates asecond delayed magnitude. The second delayed magnitude from the thirddelay circuit 814 and the cubed magnitude from the second multiplier 808are provided as inputs to the third multiplier 810. The third multiplier810 generates an output, which corresponds to the magnitude raised tothe fourth power.

The output of the third multiplier 810 is provided as an input to theseventh multiplier 830. The output of the third multiplier 810 isdelayed from the magnitude output of the magnitude computation circuit802 by the sum of the latency time of the first multiplier 806, thelatency time of the second multiplier 808, and latency time of the thirdmultiplier 810. The sixth delay circuit 820, the fifth delay circuit818, and the fourth delay circuit 816 delay samples such that Taylorseries expansion terms combined by the summing circuit 832 correspond tothe same sample.

The sixth delay circuit 820 delays the magnitude cubed output of thesecond multiplier 808 by the latency time of the third multiplier 810 totime align the magnitude cubed output with the magnitude to the fourthpower of the third multiplier 810.

The fifth delay circuit 818 delays the magnitude squared output of thefirst multiplier 806 by the sum of the latency time of the secondmultiplier 808 and the latency time of the third multiplier 810. Thefifth delay circuit 818 time aligns the magnitude squared output of thefirst multiplier 806 with the magnitude to the fourth power output ofthe third multiplier 810.

The fourth delay circuit 816 delays the magnitude output of themagnitude computation circuit 802 approximately by the sum of thelatency time of the first multiplier 806, the latency time of the secondmultiplier 808, and the latency time of the third multiplier 810. Itwill be understood by one of ordinary skill in the art that the fourthdelay circuit 816, the fifth delay circuit 818, and the sixth delaycircuit 820 can be placed in the signal path either before or after thefourth multiplier 824, the fifth multiplier 826, and the sixthmultiplier 828, respectively.

The fourth multiplier 824, the fifth multiplier 826, the sixthmultiplier 828, and the seventh multiplier 830 compute the individualterms of the Taylor series expansion. The coefficient bank 822 storesthe coefficients of the Taylor series expansion. The coefficients areapplied as inputs to the fourth multiplier 824, to the fifth multiplier826, to the sixth multiplier 828, and to the seventh multiplier 830. Theoutputs of the fourth delay circuit 816, the fifth delay circuit 818,the sixth delay circuit 820 and the third multiplier 810 are alsoapplied as inputs to the fourth multiplier 824, the fifth multiplier826, the sixth multiplier 828, and the seventh multiplier 830,respectively. In one embodiment, the latency times of the fourthmultiplier 824, the fifth multiplier 826, the sixth multiplier 828, andthe seventh multiplier 830 are approximately equal.

The outputs of the fourth multiplier 824, the fifth multiplier 826, thesixth multiplier 828, and the seventh multiplier 830 are provided asinputs to the summing circuit 832 to compute the Taylor series expansionof the preconditioning function. The output of the summing circuit 832is provided as an input to the eighth multiplier 834 and to the ninthmultiplier 836. The outputs of the first delay circuit 804 are alsoprovided as inputs to the eighth multiplier 834 and to the ninthmultiplier 836.

The first delay circuit 804 delays the in-phase component of the inputsymbol stream and the quadrature-phase component of the input symbolstream to time align the in-phase component and the quadrature-phasecomponent with the corresponding preconditioning function as provided bycomputation of the Taylor series expansion. In one embodiment, the delayof the first delay circuit 804 is approximately the sum of the latencytime of the magnitude computation circuit 802, the latency time of thefirst multiplier 806, the latency time of the second multiplier 808, thelatency time of the third multiplier 810, the latency time of theseventh multiplier 830, and the latency time of the summing circuit 832.

The preconditioning circuit 800 illustrated in FIG. 8 can be implementedin hardware or by software. For example, where the data rate isrelatively low, the preconditioning circuit 800 can be implemented bysoftware running on a general-purpose digital signal processor (DSP) ora microprocessor. In a relatively wideband application, thepreconditioning circuit 800 can be fabricated in dedicated hardwarewith, for example, a field programmable gate array (FPGA) or with anapplication specific integrated circuit (ASIC).

FIG. 9 illustrates another waveshaping circuit 900 according to oneembodiment of the present invention. The waveshaping circuit 900receives multiple input symbol streams and advantageously detects whenthe multiple input symbol streams fortuitously destructively interferewith each other such that an amount of preconditioning applied to theindividual input symbol streams can be decreased or eliminated.

In the multi-carrier waveshaping circuit 300 and the waveshaping circuit400 described earlier in connection with FIGS. 3 and 4, respectively, anindividual preconditioning circuit independently applies preconditioningto limit a relatively high signal peak in its respective input symbolstream. However, where multiple input symbol streams are eventuallycombined, such as by the first summing circuit 350 described inconnection with FIGS. 3 and 4, the multiple input symbol streams may onoccasion destructively interfere with each other. On these occasions,the preconditioning applied to relatively high signal peaks in the inputsymbol streams can be decreased or eliminated, thereby reducing oreliminating the associated injection of error energy that otherwisewould have been injected into the composite multicarrier waveform streamby the preconditioning circuits and the post-conditioning circuit.

For illustrative purposes, the waveshaping circuit 900 shown in FIG. 9processes three input symbol streams. However, it will be understood byone of ordinary skill in the art that the number of input symbol streamsprocessed by embodiments of the present invention is arbitrary. A broadrange of input symbol streams can be processed by embodiments of thepresent invention.

The illustrated waveshaping circuit 900 includes the first pulse-shapingfilter 316, the second pulse-shaping filter 318, the third pulse-shapingfilter 320, the first mixer 322, the second mixer 324, the third mixer326, the first digital NCO 328, the second digital NCO 330, the thirddigital NCO 332, and the first summing circuit 350 described earlier inconnection with FIG. 3. The waveshaping circuit 900 further includes afirst preconditioning circuit 910, a second preconditioning circuit 912,a third preconditioning circuit 914, a crest predictive weight generator916, a post-conditioning pulse generator 928, a second summing circuit930, and a delay circuit 932.

A first input symbol stream 902, a second input symbol stream 904, and athird input symbol stream 906 are applied as inputs to the firstpreconditioning circuit 910, the second preconditioning circuit 912, thethird preconditioning circuit 914, respectively, and to the crestpredictive weight generator 916. The first preconditioning circuit 910,the second preconditioning circuit 912, the third preconditioningcircuit 914, respectively, and to the crest predictive weight generator916 can be similar to the preconditioning circuits described inconnection with FIGS. 5 and 8.

A digital NCO phase information 934, a second digital NCO phaseinformation 936, and a third digital phase information 938 from thefirst digital NCO 328, the second digital NCO 330, and the third digitalNCO 332, respectively, are applied as inputs to the crest predictiveweight generator 916. The phase information allows the crest predictiveweight generator 916 to determine how the input symbol streams willcombine. The crest predictive weight generator 916 can use pulse-shapingfilters to predict how the input symbol streams will combine. In oneembodiment, the length, the latency, or both the latency and the lengthof the pulse-shaping filters of the crest predictive weight generator916 is less than the length, the latency, or both the latency and thelength of the pulse-shaping filters 316, 318, 320.

The crest predictive weight generator 916 examines the multipleinformation symbol streams and the corresponding phases of the digitalnumerical controlled oscillators to determine or to predict whether arelatively high-level signal crest will subsequently occur in thecombined signal. When the crest predictive weight generator 916 predictsthat a relatively high-amplitude signal crest will occur in the combinedsignal, the crest predictive weight generator 916 provides weight valuesto the pre-conditioning circuits that allow the preconditioning circuitsto individually process their respective input symbol streams to reducethe relatively high amplitude signal peaks. When the crest predictiveweight generator 916 predicts that destructive interference between thesymbol streams themselves will reduce or will eliminate the relativelyhigh-level signal crest, the crest predictive weight generator 916provides weight values to the preconditioning circuits that reduce ordisable the preconditioning applied by the preconditioning circuits.

The crest predictive weight generator 916 can optionally provide anadvanced crest occurrence information 940 to the post-conditioning pulsegenerator 928. The advanced crest occurrence information 940 canadvantageously be used to reduce computation latency in the waveshapingcircuit 900 by allowing the post-conditioning pulse generator 928 toinitiate early production of band-limited pulses, such as Gaussianpulses, which are applied to destructively interfere with a compositesignal output of the delay circuit 932. In other aspects, one embodimentof the post-conditioning pulse generator 928 is similar to thepost-conditioning pulse generator 348 described earlier in connectionwith FIG. 3.

One embodiment of the crest predictive weight generator 916 provides theweight value as a binary value with a first state and a second state.For example, in the first state, the crest predictive weight generator916 allows waveshaping, and in the second state, the crest predictiveweight generator 916 disables waveshaping. The crest predictive weightgenerator 916 provides the weight value or values to the preconditioningcircuits and the crest occurrence information to the post-conditioningcircuit in real time and not in non-real time. By contrast, thede-cresting control 416 described in connection with FIG. 4 can provideparameter updates to preconditioning and to post-conditioning circuitsin either real time or in non-real time. In one embodiment, awaveshaping circuit includes both the crest predictive weight generator916 and the de-cresting control 416.

The advanced crest occurrence information 940 allows the crestpredictive weight generator 916 to notify the post-conditioning pulsegenerator 928 of when the input symbol streams at least partiallydestructively interfere when combined. This allows the post-conditioningpulse generator to correspondingly decrease the magnitude of theband-limited pulse or to eliminate the band-limited pulse that wouldotherwise be applied by the post-conditioning pulse generator 928 to thecomposite signal to reduce relatively high-amplitude signal peaks.

In one embodiment, the first preconditioning circuit 910, the secondpreconditioning circuit 912, and the third preconditioning circuit 914are adapted to receive weight values 920, 922, 924 from the crestpredictive weight generator 916 and are also adapted to modify thepreconditioning according to the received weight values. In oneembodiment, the weight values 920, 922, 924 are the same for eachpreconditioning circuit and can be provided on a single signal line. Inanother embodiment, the weight values 920, 922, 924 are individuallytailored for each preconditioning circuit.

The preconditioning circuit 500 described in connection with FIG. 5 canbe modified to be used for the first preconditioning circuit 910, thesecond preconditioning circuit 912, or the third preconditioning circuit914 by allowing the applied weight value provided by the crestpredictive weight generator 916 to vary the weight applied by the weightgenerator 512. In another embodiment, the weight value from the crestpredictive weight generator 916 disables the summation of the inputsymbol stream 502 with the correction impulse by, for example, partiallydisabling the summing circuit 520, disabling the multiplier 518, or byotherwise effectively zeroing the correction impulse.

The preconditioning circuit 800 described in connection with FIG. 8 canalso be modified to be used for the first preconditioning circuit 910,the second preconditioning circuit 912, and the third preconditioningcircuit 914. For example, when the amount of preconditioning isdecreased, the weight values applied to the preconditioning circuit 800can be used to select alternative coefficients in the coefficient bank822. The weight values can also be used to decrease a magnitude of theapplied preconditioning by, for example, attenuating the output of thesumming circuit 832. Where the preconditioning is disabled, the weightvalue can be used to disable a portion of the preconditioning circuit800, such as the summing circuit 832 or the eighth multiplier 834 andthe ninth multiplier 836, to disable the preconditioning.

The waveshaping circuit 900 can further include an additional delaycircuit to compensate for computational latency in the crest predictiveweight generator 916. In one embodiment, the first preconditioningcircuit 910, the second preconditioning circuit 912, and the thirdpreconditioning circuit 914 include the additional delay circuit.

In addition to detecting when the input symbol streams destructivelyinterfere with each other so that an amount of waveshaping can bereduced or eliminated, one embodiment of the crest predictive weightgenerator 916 advantageously detects when a relatively short transitorysequence of impulses or pulses from the information source sequentiallyexhibits similar amplitude and phase levels and would otherwise giverise to a relatively large crest.

Pulse-shaping filters, such as the first pulse-shaping filter 316, thesecond pulse-shaping filter 318, and the third pulse-shaping filter 320,limit the spectral occupancy of impulse and pulse information-bearingdata streams in communication systems. A deleterious characteristic ofthese filters is that the peak to average of the pulse or impulse streamis invariably expanded during the pulse-shaping process, often by inexcess of 3 dB. These newly introduced signal crests are generallyattributed to Gibbs filter ringing effects. Ordinarily, relatively largecrests occur when a relatively short transitory sequences of impulses orpulses from the information sources sequentially exhibit similaramplitude and phase levels. These scenarios may be advantageouslypredicted by the crest predictive weight generator 916.

Upon detection of the relatively short transitory sequence of impulsesor pulses that sequentially exhibit similar amplitude and phase levels,the crest predictive weight generator 916 selects compensation with asequence of corrective vectors rather than compensation with a singlecorrective vector. This distributes the introduction of error energyover a short sequence of modulation symbols rather than to a singlesymbol. In systems that do not exploit code division multiple access(CDMA), such as Enhanced Data GSM Environment (EDGE), the distributionof the error energy is desirable because it mitigates against the impactof error energy upon the downstream receiver's detector error rate.

FIG. 10 illustrates further details of a multicarrier de-crestingcircuit 1000 according to an embodiment of the present invention. Theillustrated multicarrier de-cresting circuit 1000 does not includepre-conditioning of the input symbol streams.

The multicarrier de-cresting circuit 1000 shown in FIG. 10 includes amultiple channel circuit 1002, a de-cresting pulse generation circuit1004, and a de-cresting combiner 1006. The multiple channel circuit 1002pulse-shapes, upconverts, and combines multiple input symbol streams. Inone embodiment of the multicarrier de-cresting circuit 1000, themultiple channel circuit 1002 corresponds to a conventional circuit.Another embodiment of the multicarrier de-cresting circuit 1000 uses amultiple channel circuit described in greater detail later in connectionwith FIG. 16.

The de-cresting pulse generation circuit 1004 generates carrierwaveforms and generates post-compensation band-limited de-crestingpulses. A pulse generator control 1008 receives and inspects a compositemulticarrier signal M_(c)(t) 1010, individual subcarrier signals (orbaseband equivalents), and digital NCO waveforms. This permits the pulsegenerator control 1008 to determine the requirement for, the totalnumber of, the duration, the frequency, the amplitude and the phase ofband-limited pulses that are to be injected into the transmission datastream to reduce or to eliminate relatively high amplitude peaks in thecomposite multicarrier signal 1010. In one embodiment, the band-limitedpulses are Gaussian pulses that are provided by a bank of generalizedGaussian pulse generators that accept commands from the pulse generatorcontrol 1008 to generate a pulse of a specific duration, phase,amplitude and center frequency. Further details of the de-cresting pulsegeneration circuit 1004 are described later in connection with FIG. 15.Further details of the pulse generator control 1008 are described laterin connection with FIGS. 13A-E.

The de-cresting combiner 1006 combines the upconverted input symbolstreams with the post-compensation band-limited de-cresting pulses toremove the relatively high-level signal crests from the combined inputsymbol streams. The de-cresting combiner 1006 includes a time delaycircuit 1012. The time delay circuit 1012 delays the compositemulticarrier signal 1010 to a time-delayed composite multicarrier signal1016. The delay of the time delay circuit 1012 is matched to thecorresponding delay in the de-cresting pulse generation circuit 1004 sothat a desired amount of destructive interference can be reliablyinduced. An output of the time delay circuit 1012 is provided as aninput to a multi-input summing junction 1014, which provides ade-crested composite multicarrier signal 1018 as the linear sum of thecomposite multicarrier signal 1010, as delayed by the time delay circuit1012, and a collection of band-limited pulses. It will be understood byone of ordinary skill in the art that the band-limited pulses can beindividually applied to the multi-input summing junction 1014 or theband-limited pulses can be combined to a composite pulse stream and thenapplied to the multi-input summing junction 1014.

In one embodiment, the band-limited pulses are Gaussian pulses. Thecollection of Gaussian pulses can include zero, one, or multiple pulsesdepending on the instantaneous magnitude of the composite multicarriersignal 1010.

FIGS. 11A-E illustrate an example of the operation of the multicarrierde-cresting circuit 1000 shown in FIG. 10. With reference to FIGS.11A-E, horizontal axes 1102, 1104, 1106, 1108, 1110 indicate time. Asshown in FIGS. 11A-E, time increases to the right. FIG. 11A includes afirst waveform 1112, which corresponds to an illustrative portion of thecomposite multicarrier signal 1010. The first waveform 1112 furtherincludes a waveform crest 1114, which corresponds to a relativelyhigh-amplitude signal crest in the composite multicarrier signal 1010.Although the average power level of the composite multicarrier, signal1010 can be relatively low, the waveform crest 1114 illustrates that theinformation sources, which contribute to the input symbol streams, canoccasionally align and generate a relatively high-amplitude signal peak.For example, a signal peak that is about 10 dB above the average powerlevel can occur with a probability of 10⁻⁴. In another example, 14 dBsignal peaks can occur with a probability of 10⁻⁶.

FIG. 11B illustrates a second waveform 1116 with a pulse 1118. The pulse1118 of the second waveform 1116 corresponds to a band-limited pulse,such as a Gaussian pulse, which is generated by the de-cresting pulsegeneration circuit 1004 to destructively interfere with the relativelyhigh-amplitude signal crest in the composite multicarrier signal 1010 asillustrated by the waveform crest 1114.

FIG. 11C illustrates a third waveform 1120, which corresponds to thetime-delayed composite multicarrier signal 1016. The time delay circuit1012 delays the composite multicarrier signal 1010 to the time-delayedcomposite multicarrier signal 1016 to compensate for the computationallatency of the de-cresting pulse generation circuit 1004. This alignmentis shown in FIGS. 11B and 11C by the alignment of the delayed signalcrest 1122 with the pulse 1118.

The band-limited pulse destructively interferes with the relatively highsignal peak in the time-delayed composite multicarrier signal 1016. FIG.11D illustrates a fourth waveform 1124, which corresponds to the outputof the multi-input summing junction 1014. The fourth waveform 1124 isthus the linear superposition of the second waveform 1116 and the thirdwaveform 1120. In the fourth waveform 1124, a compensated portion 1126is substantially devoid of the waveform crest 1114 by the destructiveinterference induced by the band-limited pulse. FIG. 11E superimposesthe second waveform 1116, the third waveform 1120, and the fourthwaveform 1124.

FIGS. 12A-C illustrate a complementary frequency domain analysis of themulticarrier de-cresting circuit that uses only a single Gaussian pulseto de-crest a composite waveform. FIG. 12A illustrates an example of abasic power spectral density plot (PSD) of a composite single carriersignal 1202 and a PSD plot of a single Gaussian pulse 1204. FIG. 12Aalso illustrates a resulting output signal power spectral density 1206when the composite single carrier signal 1202 and the single Gaussianpulse 1204 are linearly combined. In one embodiment, the multicarrierde-cresting circuit 1000 expands the PSD only when the Gaussian pulse'scharacteristics expand the signal energy beyond the basic frequencyallocation. Thus, the bandwidth expansion of the combined signal isreadily controlled by controlling the characteristics of the de-crestingpulse generation circuit 1004 configured to generate a single Gaussianpulse.

FIG. 12B also illustrates the applicability of a generating a singleGaussian pulse to reduce a magnitude of a signal crest in a multicarrierapplication. A trace 1208 corresponds to a basic PSD plot correspondingto a multicarrier signal crest. A trace 1210 corresponds to a PSD plotof the single Gaussian pulse. A trace 1212 illustrates a composite PSDof the combination of the multicarrier signal crest with the singleGaussian pulse.

FIG. 12C illustrates a disadvantage of generating a single Gaussianpulse to reduce the magnitude of a signal crest in a multicarriersignal. In the example shown in FIG. 12C, one of the channel streams isdropped either temporarily or permanently from the compositemulticarrier signal 1010. A trace 1214 corresponds to a basic PSD of themulticarrier signal crest with a channel stream dropped. A trace 1216corresponds to a PSD plot of the single Gaussian pulse. A trace 1218illustrates a composite PSD of the combination of the single Gaussianpulse and the multicarrier signal crest with the channel stream dropped.As shown in FIG. 12C, energy from the Gaussian pulse increases theresidual energy level within the unoccupied channel allocation. Theincrease in residual energy in the unoccupied channel is relativelyundesirable in a commercial application.

Embodiments of the invention, such as the multicarrier de-crestingcircuit 1000 described in connection with FIG. 10, advantageouslyovercome the undesirable polluting of unoccupied channel allocations byinjecting multiple band-limited pulses from multiple pulse generators.In one embodiment, the multiple band-limited pulses are Gaussian pulses.The generation of multiple band-limited pulses allows the pulsegenerator control to determine the PSD content in each of the allocatedchannels and advantageously insert Gaussian pulse energy only intooccupied channels to counteract the signal peak. This advantageouslyprevents the injection of Gaussian pulse energy to unoccupied channelallocation.

Further, one embodiment of the pulse generator control 1008 is providedwith the individual amplitude levels for each baseband channel'scontribution to the overall composite signal's peak, so that the pulsegenerator control 1008 can weigh the amplitude of each Gaussian pulseaccording to the contribution to the peak in the composite multicarriersignal 1010.

FIGS. 13A-E illustrate the operation of the pulse generator control 1008described in connection with FIG. 10. The pulse generator control 1008advantageously provides multiple band-limited pulses, such as Gaussianpulses, that destructively interfere with the signal crests in thecomposite multicarrier signal 1010. With reference to FIGS. 13A-E,horizontal axes 1302, 1304, 1306, 1308, 1310 indicate time. As shown inFIGS. 13A-E, time increases to the right.

FIG. 13A includes a first waveform 1312, which corresponds to a portionof the composite multicarrier signal 1010. The first waveform 1312further includes a waveform crest 1314, which corresponds to arelatively high-amplitude signal crest in the composite multicarriersignal 1010. The first waveform 1312 and the waveform crest 1314 aresimilar to the first waveform 1112 and the waveform crest 1114 describedin connection with FIG. 11A.

FIG. 13B illustrates a second waveform 1316 that includes cancellationpulses 1318, 1320 that are generated from a family of band-limitedpulses 1322, 1324, 1326, 1328, 1330, such as Gaussian pulses. Incontrast to a single destructive pulse, such as the pulse 1118 describedearlier in connection with FIG. 11B, the cancellation pulses 1318, 1320in the second waveform 1316 include multiple cancellation pulses. Thepulses in the family of band-limited pulses 1322, 1324, 1326, 1328, 1330are selected to be centered at the corresponding active channelfrequencies. The cancellation pulses 1318, 1320 of the second waveform1316 are generated by the de-cresting pulse generation circuit 1004 todestructively interfere with the relatively high-amplitude signal crestin the composite multicarrier signal 1010 as illustrated by the waveformcrest 1314.

FIG. 13C illustrates a third waveform 1332, which corresponds to thetime-delayed composite multicarrier signal 1016. The time delay circuit1012 delays the composite multicarrier signal 1010 to the time-delayedcomposite multicarrier signal 1016 to compensate for the computationallatency of the de-cresting pulse generation circuit 1004. This alignmentis shown in FIGS. 13B and 13C by the alignment of a delayed signal crest1334 with the cancellation pulses 1318, 1320.

The cancellation pulses 1318, 1320 destructively interfere with therelatively high signal peak in the time-delayed composite multicarriersignal 1016. FIG. 13D illustrates a fourth waveform 1336, whichcorresponds to the output of the multi-input summing junction 1014. Thefourth waveform 1336 is thus the linear superposition of the secondwaveform 1316 and the third waveform 1332. In the fourth waveform 1336,a compensated portion 1338 is substantially devoid of the waveform crest1314 by the destructive interference induced by the band-limited pulse.FIG. 13E superimposes the second waveform 1316, the third waveform 1332,and the fourth waveform 1336.

FIGS. 14A and 14B illustrate the results of a complementary frequencydomain analysis of the multicarrier de-cresting circuit 1000. Withreference to FIG. 14A, a trace 1402 is a basic PSD plot of the compositemulticarrier signal 1010, which is provided as an input to thede-cresting pulse generation circuit 1004. A trace 1404 is a PSD plot ofthe multiple Gaussian pulses, which are the outputs of the de-crestingpulse generation circuit 1004. A trace 1406 is a PSD plot of thede-crested composite multicarrier signal 1018 of the multi-input summingjunction 1014, which combines the time-delayed composite multicarriersignal 1016 with the multiple Gaussian pulses. The trace 1406illustrates that the PSD bandwidth expansion of the de-crested compositemulticarrier signal 1018 can be relatively readily controlled bymanaging the PSD of the corresponding multiple Gaussian pulses from thede-cresting pulse generation circuit 1004.

With reference to FIG. 14B, a trace 1408 is a PSD plot of the compositemulticarrier signal 1010, where the composite multicarrier signal 1010includes a non-utilized channel allocation. Advantageously, embodimentsof the invention can inject multiple Gaussian pulses to destructivelyinterfere with signal peaks at the utilized channel allocations, therebypreventing the expansion or pollution of the frequency spectrum. A trace1410 is a PSD plot of multiple Gaussian pulses, which correspond tooutput of the de-cresting pulse generation circuit 1004. In oneembodiment, each of the multiple Gaussian pulses generated by thede-cresting pulse generator is substantially band-limited to itscorresponding channel. A trace 1412 is a PSD plot of the de-crestedcomposite multicarrier signal 1018 of the multi-input summing junction1014, which combines the time-delayed composite multicarrier signal 1016with the multiple Gaussian pulses. In contrast to the injection of asingle Gaussian pulse de-crest the composite multicarrier signal 1010,which is illustrated in FIG. 12C, the injection of multiple Gaussianpulses corresponding only to allocated channels is advantageouslyrelatively free from spectral pollution.

FIG. 15 illustrates one embodiment of the de-cresting pulse generationcircuit 1004. The de-cresting pulse generation circuit 1004advantageously provides multiple band-limited pulses to de-crest thecomposite multicarrier signal 1010 with relatively little pollution ofthe frequency spectrum.

The illustrated de-cresting pulse generation circuit 1004 includes thepulse generator control 1008 and a pulse generator 1502. The pulsegenerator control 1008 shown in FIG. 15 further includes a comparator1504, a weight generator 1506, and an impulse generator 1508.

The composite multicarrier signal 1010 is provided as an input to thecomparator 1504. In addition, the comparator 1504 receives channelinputs from the pulse shaping filters and phase information from digitalNCO sources. This information enables the comparator 1504 to determinewhether to apply single or multiple cancellation pulses to de-crest thecomposite multicarrier signal 1010 or the time-delayed compositemulticarrier signal 1016. In one embodiment, the comparator 1504compares these signals to reference information of the intrinsicwaveform. The reference information can include the average, the peak,and other pertinent signal statistics to determine whether to applycancellation pulses to de-crest the composite multicarrier signal 1010.

When the comparator 1504 has determined that a cancellation pulse or agroup of cancellation pulses will be applied, the comparator 1504calculates a duration for a cancellation pulse and instructs the impulsegenerator 1508 to provide a sequence of impulses to the pulse generator1502.

The weight generator 1506 provides weight values to the pulse generator1502. The weight values are used by the pulse generator 1502 to vary anamount of a band-limited de-cresting pulse injected into a channelaccording to the weight value corresponding to the channel.

In one embodiment, the weight generator 1506 calculates a relativemagnitude and phase for each channel's contribution to the crest in thecomposite multicarrier signal 1010 and provides weight values to thepulse generator 1502 so that each channel suffers an approximately equaldegradation in signal quality. The weight values generated by the weightgenerator 1506 can advantageously be set at a zero weight for inactivechannels and a relatively high weight for relatively high-powerchannels. The weight values can correspond to positive values, tonegative values, to zero, and to complex values. This allows the errorvector magnitude (EVM) to be approximately equal for all activechannels, while simultaneously eliminating or reducing signal crests.

In another embodiment, a single active channel is randomly selected forintroduction of a stronger correction pulse. This lowers aggregate errorrates, but increases the severity of the errors.

The pulse generator 1502 includes a group of multipliers 1510, a groupof filters 1512, and a summing circuit 1514. It will be understood byone of ordinary skill in the art that the waveshaping circuits andsub-circuits disclosed herein can be configured to process an arbitraryor “N” number of channels. In addition, although the pulse generator1502 can include processing capability for several channels, it will beunderstood by one of ordinary skill in the art that some applicationswill not utilize all of the processing capability.

The group of multipliers 1510 in the illustrated pulse generator 1502can include “N” multipliers. A first multiplier 1516 multiplies theimpulses from the pulse generator 1502 with the weight value from theweight generator 1506 that corresponds to a first channel. A secondmultiplier 1518 similarly multiplies the impulses from the pulsegenerator 1502 with the weight value from the weight generator 1506 thatcorresponds to a second channel.

The group of filters 1512 in the illustrated pulse generator 1502 caninclude “N” passband filters. A first passband filter 1520 generatesband-limited pulses in response to receiving impulses from the firstmultiplier 1516. The band-limited pulses from the first passband filter1520 are centered at approximately the first channel's frequency band orallocation. In one embodiment, the first passband filter 1520 is aGaussian passband finite impulse response (FIR) filter.

A second passband filter 1522 similarly generates band-limited pulses inresponse to receiving impulses from the second multiplier 1518. Theband-limited pulses from the second passband filter 1522 are centered atapproximately the second channel's frequency band or allocation. In oneembodiment, the second passband filter 1522 is a Gaussian passband FIRfilter. Preferably, all passband filters in the group of filters 1512are FIR filters so that the outputs of the passband filters are phasealigned.

The summing circuit 1514 combines the outputs of the first passbandfilter 1520, the second passband filter 1522, and other passbandfilters, as applicable, in the group of filters 1512. The output of thesumming circuit 1514 is a composite stream of Gaussian pulses, which isthen applied to the multi-input summing junction 1014 to reduce or toeliminate relatively high amplitude signal crests. In anotherembodiment, the individual outputs of the passband filters in the groupof filters 1512 are applied directly the multi-input summing junction1014.

FIG. 16 illustrates a multiple channel circuit 1600 according to anembodiment of the present invention. The multiple channel circuit 1600advantageously reduces the likelihood of the occurrences of signalcrests in composite waveforms, and can be used to decrease a frequencyof application of waveshaping. It will be understood by one of ordinaryskill in the art that the number of channels pulse shaped and combinedby the multiple channel circuit 1600 can be arbitrarily large.

The multiple channel circuit 1600 includes fractional delays, whichstagger the input symbol streams relative to each other by fractions ofa symbol period. In one embodiment, the delay offset from one symbolstream to another is determined by allocating the symbol period over thenumber of active symbol streams. For example, where “x” corresponds to asymbol period and there are four input symbol streams, a first symbolstream can have 0 delay, a second input symbol stream can have 0.25xdelay, a third input symbol stream can have 0.50x delay, and a fourthinput symbol stream can have 0.75x delay.

The illustrated embodiment of the multiple channel circuit 1600implements the fractional delay to the data streams before the pulseshaping filters. In one example, “N,” or the number of active symbolstreams, corresponds to 4. In the multiple channel circuit 1600, a firstinput symbol stream 1602 is applied as an input directly to a firstpulse-shaping filter 1604 without fractional delay. In anotherembodiment, the data stream associated with the first input symbolstream 1602 includes a fractional delay.

A second input symbol stream 1606 is provided as an input to a firstfractional delay circuit 1608, which delays the second input symbolstream 1606 relative to the first input symbol stream 1602 by a firstfraction of a symbol period, such as 0.25 of the symbol period. A thirdinput symbol stream 1612 is provided as an input to a second fractionaldelay circuit 1614, which delays the third input symbol stream 1612relative to the first input symbol stream 1602 by a second fraction ofthe symbol period, such as 0.50 of the symbol period. A fourth inputsymbol stream 1618 is applied to a third fractional delay circuit 1620,which delays the fourth input symbol stream 1618 by a third fraction ofa symbol period, such as 0.75 of the symbol period.

The staggered symbol streams are mixed by their respective mixercircuits 1624, 1626, 1628, 1630 and combined by a summing circuit 1632.The staggering of the symbol streams reduces the probability ofoccurrence of signal crests in the resulting composite waveform 1634because the staggering displaces each channel's individual signal crestfrom another channel's signal crest as a function of time. Thisdecreases the probability of a mutual alignment in amplitude and phasein the composite waveform 1634.

However, it will be understood by one of ordinary skill in the art thatthe fractional delay can be applied elsewhere, such as embedded directlywithin a pulse-shaping filter, applied post pulse-shaping, and the like.In one embodiment, the amount of the fractional delay for each symbolstream is fixed in hardware. In another embodiment, the fractionaldelays can be selected or programmed by, for example, firmware.

Some systems that are susceptible to relatively high-amplitude signalpeaks or crests are incompatible with techniques that modify theamplitude of the underlying signals to reduce or to eliminate therelatively high-amplitude signal peaks in a composite multicarriersignal. One example of such a system is an EDGE system, whereintroduction of amplitude modulating pulses such as band-limitedGaussian pulses is undesirable and may not be permissible.

FIG. 17 illustrates a phase-modulating waveshaping circuit 1700according to an embodiment of the present invention. Advantageously, thephase-modulating waveshaping circuit 1700 reduces or eliminatesrelatively high-amplitude signal crests in composite multi-carriersignals without modulation of the amplitude of the underlying signals.Rather than sum a composite multicarrier signal with band-limited pulsesto de-crest the composite multicarrier signal as described in connectionwith FIG. 10, the phase-modulating waveshaping circuit 1700 modulatesthe phases of the input symbol streams to reduce or to eliminaterelatively high signal crests in the resulting composite multicarriersignal. It will be understood by one of ordinary skill in the art thatthe phase-modulating waveshaping circuit 1700 can be configured toprocess an arbitrary or “N” number of channels.

The phase-modulating waveshaping circuit 1700 includes a multiplechannel circuit 1702, a de-cresting combiner 1704, digital NCOs 1706,and a pulse phase modulation circuit 1708. The multiple channel circuit1702 receives the input symbol streams, pulse shapes and upconverts theinput symbol streams. The pulse shaped and upconverted input streams areprovided as inputs to the de-cresting combiner 1704 and to a pulse phasemodulator control 1710 of the pulse phase modulation circuit 1708.

One embodiment of the pulse phase modulation circuit 1708 issubstantially the same as the de-cresting pulse generation circuit 1004described in connection with FIGS. 10 and 15. However, rather thansumming the composite multicarrier signal with the generatedband-limited pulses, the band-limited pulses are used to phase modulatethe upconverted symbol streams. As such, the pulse phase modulatorcontrol 1710 corresponds to the pulse generator control 1008. The pulsephase modulator control 1710 predicts whether the current modulationstreams and digital NCO phase combinations will constructively interferewith each other and result in a composite waveform crest. Where a crestis predicted, the Gaussian pulse phase modulators are engaged torelatively slowly modulate the individual channel phases to prevent orto reduce a signal crest in the composite waveform.

A Gaussian pulse phase modulator, such as a first Gaussian pulse phasemodulator 1712 corresponds to a Gaussian pulse generator, such as afirst Gaussian pulse generator 1020. Again, the corresponding Gaussianpulses gp₁(t), gp₂(t), and so forth, generated by the Gaussian pulsephase modulators of the pulse phase modulation circuit 1708 areband-limited to their corresponding input symbol stream's allocatedchannel.

The de-cresting combiner 1704 includes multiple delay circuits 1714,1716, 1718, 1720, which align the upconverted symbol streams from themultiple channel circuit 1702 with the Gaussian pulses from the pulsephase modulation circuit 1708. The de-cresting combiner 1704 furtherincludes phase modulators 1722, 1724, 1726, 1728, which phase modulatetheir respective upconverted input symbol streams in accordance with therespective Gaussian pulse from the pulse phase modulation circuit 1708.A summing circuit 1730 combines the outputs of the phase modulators1722, 1724, 1726, 1728 and provides a de-crested composite multicarriersignal 1732 as an output.

The skilled practitioner will recognize that care should be taken toensure that the rate of change of phase due to this correction processdoes not exceed the capability of the downstream receivers to trackeffective channel phase variations.

One embodiment of the present invention further uses a pulse generatorcontrol or a pulse phase modulator control that is already used tode-crest or to waveshape composite signals to continually monitor and toreport the amplitude and phase information of each individual basebandchannel. This information can be readily utilized to extract the averageand peak power levels of individual channels. In addition, the presenceof active or dormant channels can be readily ascertained. Thisinformation is extremely useful for external subsystems in a range ofcommunications applications.

In one embodiment, a waveshaping circuit includes a communications port,such as a serial communications port or a parallel communications portthat enables this information to be transmitted to external devices. Inanother embodiment, the collected information is stored in a memorystructure, which is accessed by multiple external devices requiring suchinformation. The information can be ported to an amplifier linearizationchip such as the PM7800 PALADIN product from PMCS.

One embodiment of the waveshaping circuit is implemented in dedicatedhardware such as a field programmable gate array (FPGA) or dedicatedsilicon in an application specific integrated circuit (ASIC). In arelatively low data rate application, a general purpose digital signalprocessor (DSP), such as a TMS320C60 from Texas Instruments Incorporatedor a SHARC processor from Analog Devices, Inc., performs the waveshapingsignal processing.

A conventional microprocessor/microcontroller or general purpose DSP caninterface to a waveshaping circuit to adaptively control the waveshapingprocess. For example, a de-cresting control can operate in non-realtime, and a general purpose DSP or microprocessor such as aTMS320C54/TMS320C60/TMS320C40/ARM 7 or Motorola 68000 device can be usedfor control. Preferably, the DSP or microprocessor includes non-volatileROM for both program storage and factory installed default parameters.Both ROM and Flash ROM are relatively well suited for this purpose. Aswith most DSP or microprocessor designs, a proportional amount of RAM isused for general-purpose program execution. In one embodiment, arelatively low speed portion of the waveshaping circuit implemented witha DSP or a microprocessor core and a relatively high speed portion ofthe waveshaping circuit implemented in an ASIC or an FPGA is integratedonto a single ASIC chip with an appropriate amount of RAM and ROM.Examples of licensable cores include the ARM7 from Advanced RISCMachines, Ltd., the Teak from DSP Group Inc., the Oak from DSP GroupInc., and the ARC from ARC Cores.

Various embodiments of the present invention have been described above.Although this invention has been described with reference to thesespecific embodiments, the descriptions are intended to be illustrativeof the invention and are not intended to be limiting. Variousmodifications and applications may occur to those skilled in the artwithout departing from the true spirit and scope of the invention asdefined in the appended claims.

1. A pulse-shaping circuit that reduces a probability of an alignment inamplitude and phase of similar symbols in a plurality of input symbolstreams including at least a first input symbol stream and a secondinput symbol stream, where the plurality of input symbol streams areeventually upconverted and combined to a composite data stream such thata reduction in the probability of the alignment reduces a probability ofa large signal crest in the composite data stream, the pulse-shapingcircuit comprising: a plurality of pulse-shaping filters adapted topulse-shape the plurality of input symbol streams to form acorresponding plurality of baseband streams; a plurality of multipliersadapted to receive a plurality of input carrier streams and to upconvertthe plurality of baseband streams to a plurality of upconverted streams;a summing circuit that combines the upconverted streams to the compositesignal; and a delay circuit in at least a first data path, where thefirst data path is a path from an input symbol stream to the compositedata stream, where the delay circuit delays data in the first data pathby a fraction of a symbol period relative to data in a second data path.2. The pulse-shaping circuit as defined in claim 1, wherein the at leastfirst data path comprises a plurality of data paths with delay circuits,wherein the delay circuits delay data in their respective data pathsrelative to data in the second data path with delays distributedapproximately evenly through the symbol period.
 3. The pulse-shapingcircuit as defined in claim 1, wherein the delay circuit in the firstdata path is configured to delay data between the first input symbolstream and the first pulse shaped filter.
 4. A method of digitallyreducing a probability of an alignment in amplitude and phase of similarsymbols in a plurality of data streams, the method comprising: receivingthe plurality of data streams, where the plurality of data streamsinclude at least a first data stream and a second data stream; delayingthe second data stream relative to the first data stream by a fractionof a symbol period prior to the eventual combining of the plurality ofdata streams; and pulse-shaping, upconverting, and combining theplurality of data streams to a composite data stream such that areduction in the probability of the alignment reduces a probability oflarge signal crest in the composite data stream.
 5. The method asdefined in claim 4, wherein the plurality of data streams includes thefirst data stream and other data streams, wherein the fraction of thesymbol period from which the other data streams are delayed relative tothe first data stream substantially evenly through the symbol period. 6.The method as defined in claim 4, wherein the plurality of data streamsincludes “N” data streams, wherein the fraction of the symbol periodfrom a data stream other than the first data stream is delayed relativeto the first data stream by about a fraction of time equal to a multipleof the symbol period divided by N, so that the data streams other thanthe first data stream are substantially evenly delayed through thesymbol period.
 7. An apparatus for digitally reducing a probability ofan alignment in amplitude and phase of similar symbols in a plurality ofdata streams, the apparatus comprising: means for receiving theplurality of data streams, where the plurality of data streams includeat least a first data stream and a second data stream; means fordelaying the second data stream relative to the first data stream by afraction of a symbol period prior to the eventual combining of theplurality of data streams; and means for pulse-shaping, upconverting,and combining the plurality of data streams to a composite data streamsuch that a reduction in the probability of the alignment reduces aprobability of large signal crest in the composite data stream.
 8. Theapparatus as defined in claim 7, wherein the plurality of data streamsincludes the first data stream and other data streams, wherein thefraction of the symbol period from which the other data streams aredelayed relative to the first data stream substantially evenly throughthe symbol period.
 9. The apparatus as defined in claim 7, wherein theplurality of data streams includes “N” data streams, wherein thefraction of the symbol period from a data stream other than the firstdata stream is delayed relative to the first data stream by about afraction of time equal to a multiple of the symbol period divided by N,so that the data streams other than the first data stream aresubstantially evenly delayed through the symbol period.